1. Field of the Invention
The present invention relates to a switching power supply circuit equipped to various types of electronic equipment as a power source.
2. Description of the Related Art
There has been widely known a switching power supply circuit using a switching converter of such a type as a fly-back converter or a forward converter. These switching converters are restricted in reduction of switching noises because the switching operation waveform thereof is a rectangular waveform. Further, it has been found that restrictions are imposed on improvements of the power transform efficiency from the viewpoint of the performance characteristics.
Therefore, various types of switching power supply circuits each based on a resonance type converter were previously proposed by the applicant of this application. The resonance type converter can easily achieve a high power transform efficiency, and also it can reduce the noises because the switching operation waveform is a sine wave. Further, there is a merit that it can be constructed by a relatively small number of parts.
FIG. 7 is a circuit diagram showing a conventional switching power supply circuit, which can be constructed on the basis of the invention previously-proposed by the applicant of this application. As the basic construction of the power supply circuit shown in FIG. 7, it is equipped with a voltage resonance type converter as a primary switching converter.
In the power supply circuit shown in FIG. 7, a rectified smoothened voltage Ei corresponding to the level which is once as high as an alternating input voltage VAC is generated from a commercial alternating power source (alternating input voltage VAC) by a bridge rectifying circuit Di and a smoothing capacitor Ci.
At the primary side of the power supply circuit thus constructed, a self-exciting type construction is shown as a voltage resonance type converter circuit for performing a single-end operation by a single-stone switching element Q1. In this case, a bipolar transistor (BJT; junction type transistor) having high resistance to voltage is adopted for the switching element Q1.
The base of the switching element Q1 is connected to the positive polarity side of the smoothing capacitor Ci (rectified smoothened voltage Ei) through a starting resistor (RS), and the base current at the starting time is achieved from the rectifying and smoothing line.
A drive winding NB comprising one turn 1T of winding at the primary side of the insulating converter transformer PIT, and a series resonance circuit for self-exciting driving operation which comprises a series circuit of an inductor LB, a resonance capacitor CB and a base current limiting resistor RB are connected across the base of the switching element Q1 and the earth at the primary side. A switching frequency fs for switching on/off the switching element Q1 is generated by the self-exciting circuit.
A route for clamp current flowing when the switching element Q1 is in off-state is formed by a clamp diode DD1 inserted between the base of the switching element Q1 and the negative polarity (the earth at the primary side) of the smoothing capacitor Ci. Further, the collector of the switching element Q1 is connected to the winding-start edge portion of the primary winding N1 of the insulating converter transformer PIT, and the emitter thereof is connected to the earth.
A parallel resonance capacitor Cr is connected between the collector and emitter of the switching element Q1 in parallel to the switching element Q1. In this case, the primary parallel resonance circuit of the voltage resonance type converter is also formed by the capacitance of the parallel resonance capacitor cr itself and a leakage inductance L1 at the primary winding N1 side of the insulating converter transformer PIT.
The insulating converter transformer PIT is provided to transmit the switching output of the switching converter achieved at the primary side to the secondary side.
The insulating converter transformer PIT is provided with an EE type core comprising ferrite E type cores CR1, CR2 as shown in FIG. 8. In the insulating converter transformer PIT, divided bobbins B are used, and the primary winding N1 and the secondary winding N2 both of which are litz wires are wounded around the divided areas as shown in FIG. 8. Here, the primary winding N1 and the secondary winding N2 are wound in the same winding direction.
A gap G is formed for a center magnetic leg of the EE type core as shown in FIG. 8. The leakage inductance in the insulating converter transformer PIT is determined by the gap length of the gap G, and loose coupling based on a required coupling coefficient is achieved. The coupling coefficient k at this time is set to k≈0.85 so that the loose coupling state is achieved, and thus the saturation state is hardly achieved. The gap G can be formed by making the center magnetic leg of the E-type cores CR1, CR2 shorter than two outer magnetic legs, and the gap length in this case is set to about 1 mm.
For the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2, the operation of the insulating converter transformer PIT may be selectively set to a +M operation mode (additive polarity mode: forward operation) or a xe2x88x92M operation mode (subtractive polarity mode: fly-back operation) in accordance with the connection relationship between the polarity (winding direction) of the primary winding N1, the secondary winding N2 and the rectifying diode D0.
For example, assuming that the polarities (winding directions) of the primary winding N1 and the secondary winding N2 are the same, the mutual inductance is set to +M if the circuit is equivalent to the circuit shown in FIG. 9A, and the mutual inductance is set to xe2x88x92M if the circuit is equivalent to the circuit shown in FIG. 9B.
As shown in FIG. 7, the winding-start edge portion of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the main switching element Q1, and the winding-end edge portion is connected to the line of the rectified smoothened voltage Ei.
Further, the winding-start edge portion of the secondary winding N2 is connected to the earth at the secondary side, and the winding-end edge portion is connected to the positive-polarity terminal of the smoothing capacitor C01 through the rectifying diode D01.
In such a connection style, the additive polarity connection is carried out between the primary winding N1 and the secondary winding N2 of the insulating converter transformer PIT, and this corresponds to the equivalent circuit shown in FIG. 9A.
The switching output of the main switching element Q1 forming the primary voltage resonance type converter is transmitted to the primary winding N1 of the insulating converter transformer PIT having the above construction, and further transmitted to the secondary winding N2 while it is excited.
In this case, at the secondary side of the insulating converter transformer PIT, the secondary parallel resonance capacitor C2 is connected to the secondary winding N2 in parallel as shown in the figure, so that the secondary parallel resonance circuit is formed together with the leakage inductance L2 of the secondary winding N2.
A half-wave rectifying circuit comprising the rectifying diode D01 and the smoothing capacitor C01 is connected to the secondary parallel resonance circuit in the connection style shown in the figure, thereby outputting the secondary DC output voltage E01.
In the power supply circuit thus constructed, the primary side is equipped with the parallel resonance circuit for setting the switching operation to the voltage resonance type, and the secondary side is equipped with the parallel resonance circuit for achieving the voltage resonance operation. In this specification, the switching converter that operates while it is equipped with the resonance circuits at the primary and secondary sides is referred to as xe2x80x9ccomposite resonance type switching converterxe2x80x9d.
Further, in the power supply circuit, an active clamp circuit 20 is equipped to the secondary side.
That is, as the secondary active clamp circuit 20 are provided an auxiliary switching element Q2 of MOS-FET, a clamp capacitor C3, and a clamp diode DD2 of a body diode. Further, a drive winding Ng1, a capacitor Cg1 and a resistor Rg1 are equipped as a driving circuit system for driving the auxiliary switching element Q2.
A clamp diode DD2 is connected in parallel between the drain and source of the auxiliary switching terminal Q2. As a connection style, the anode of the clamp diode DD2 is connected to the source, and the cathode is connected to the drain.
Further, the drain of the auxiliary switching element Q2 is connected to the connection point between the winding-end edge portion of the secondary winding N2 and the anode of the rectifying diode D01 through the clamp capacitor C3. The source of the auxiliary switching element Q2 is connected to the secondary earth.
Accordingly, the secondary active clamp circuit 20 is constructed by connecting the clamp capacitor C3 to the parallel connection circuit of the auxiliary switching element Q3, the clamp diode DD2 in series. The circuit thus formed is further connected to the secondary parallel resonance circuit in parallel.
Further, as the driving circuit system of the auxiliary switching element Q2, the series connection circuit of capacitor Cg1-resistor Rg1-drive winding Ng1 is connected to the gate of the auxiliary switching element Q2 as shown in the figure. The series connection circuit forms a self-exciting type driving circuit for the auxiliary switching element Q2. That is, a signal voltage is applied from the self-exciting type driving circuit to the gate of the switching element Q2 to carry out the switching operation.
In this case, the driving winding Ng1 is formed at the winding-start edge portion side of the secondary winding N2, and the number of turns is set to 1T (turn) , for example.
Accordingly, a voltage excited by an alternating voltage achieved at the primary winding N1 occurs at the drive winding Ng1. In this case, voltages having the opposite polarities are achieved at the secondary winding N2 and the drive winding Ng1 from the viewpoint of the relationship of the winding direction.
The switching operation of the auxiliary switching element Q2 is subjected to PWM control by the control circuit 1 equipped at the secondary side.
That is, the secondary DC output voltage E01 is supplied to the control circuit 1, and the control circuit 1 applies the DC control voltage corresponding to the secondary DC output voltage E01 to the gate of the auxiliary switching element Q2 to control the conduction angle of the auxiliary switching element Q2, whereby the stabilization of the DC output voltage E01 to the variation of the alternating input voltage VAC and the load power Po is carried out.
In the power supply circuit shown in FIG. 7, the winding directions of the primary winding N1 and the secondary winding N2 are the same as shown by the structure of the insulating converter transformer PIT of FIG. 8. Accordingly, magnetomotive force is generated at the primary winding N1 by primary winding current I1 flowing through the primary winding Ni. Likewise, magnetomotive force is generated at the secondary winding N2 by secondary winding current I2 flowing through the secondary winding N2, whereby a primary magnetic flux xcfx861 occurs at the primary side while a secondary magnetic flux xcfx862 occurs at the secondary side as shown in FIG. 8.
As described above, the primary winding N1 and the secondary winding N2 in the circuit of FIG. 7 are connected to each other with additive polarity, so that the primary magnetic flux xcfx861 and the secondary magnetic flux xcfx862 work to be added with each other. Accordingly, a magnetic flux represented by xcfx861+xcfx862 occurs at the center magnetic leg of the insulating converter transformer PIT.
That is, the primary winding N1 and the secondary winding N2 have the same winding direction and satisfy the additive polarity connection, so that a relatively large magnetic flux comprising the mixture of the primary magnetic flux xcfx861 and the secondary magnetic flux xcfx862 occurs at the center magnetic leg.
Here, if no gap is formed at the center magnetic leg of the core of the insulating converter transformer PIT (gap length=0), the magnetic flux enters a saturation area of the magnetization curve of the ferrite core under the condition that the load power Po=about 100W, for example. In the specification, the xe2x80x9csaturationxe2x80x9d situation means the state that the magnetic flux enters such a saturation area of the magnetization curve.
Accordingly, the inductance of the core is sharply reduced, and the main switching element Q1 of BJT may be broken with high probability.
Therefore, the insulating converter transformer PIT is designed so that the loose coupling state based on a required coupling coefficient can be achieved by forming the gap G as shown in FIG. 8, whereby no saturation occurs.
In order to avoid the phenomenon described above and satisfy the regulation range in the case of the power supply circuit having the construction shown in FIG. 7, it is required to manage the gap length of the gap G formed in the insulating converter transformer PIT with the precision of 1 mmxc2x10.1 mm.
In order to satisfy the precision of the gap length described above, it is required to polish the center magnetic leg of each of the E-type core CR1, Cr2 and carry out the manufacturing management with the precision of 0.5 mmxc2x10.05 mm. Accordingly, the manufacturing time is increased because a work of polishing the center magnetic leg of the E type core with high precision is needed, and it is difficult to perform the product management because there is considered such a case that insulating converter transformers which have the same E type cores, but are different in gap length are produced. That is, necessity of forming a gap causes the manufacturing efficiency to be lowered.
When the gap G is formed in the insulating converter transformer PIT, a leakage magnetic flux called as a fringe magnetic flux occurs in the neighborhood of the gap G, so that an eddy current loss occurs at the primary winding N1 and the second winding N2 corresponding to litz wires, and local heat occurs. This heat is transferred to wires under a low temperature, and the temperature of the windings themselves is increased. Accordingly, it has been found that a power loss called as a copper loss is increased and the power transform efficiency is lowered.
Particularly, in the circuit shown in FIG. 7, the high-frequency current amount of the primary winding current I1 flowing in the primary winding N1 and the secondary winding current I2 flowing in the secondary winding N2 is large, so that the heat due to the DC resistance as the litz wire and the eddy current loss in the primary-winding current I1 and the second winding N2 is remarkable.
Further, in the circuit shown in FIG. 7, there occurs such a problem that when the level of the alternating input voltage VAC under a heavy load condition is lowered to the level of about 75 V to 85 V in AC 100 system, an abnormal operation period which is not the ZVS (Zero Voltage Switching) operation occurs as the operation of the primary main switch element Q1. If such a phenomenon lasts, the main switch element Q1 is heated, and it may be broken in a short time.
Therefore, in view of the foregoing problem, there is provided a switching power supply circuit comprising: switching means formed to have a main switching element for intermittently outputting a DC input voltage; a primary parallel resonance capacitor provided so as to form a primary parallel resonance circuit for making the operation of the switching means a voltage resonance type; an insulating converter transformer having a structure that a required coupling coefficient to establish the loose coupling between the primary side and the secondary side is achieved, the insulating converter transformer transmitting the output of the switching means achieved at the primary side to the secondary side; a secondary resonance circuit formed by connecting a secondary resonance capacitor to a secondary winding of the insulating converter transformer; DC output voltage generating means that receives an alternating voltage achieved at the second winding of said insulating converter transformer to carry out a rectifying operation, thereby achieving a secondary DC output voltage; secondary active clamp means that is formed in parallel to said secondary resonance capacitor so as to have a series connection circuit comprising a clamp capacitor and a secondary auxiliary switching element; and voltage stabilizing means for applying a DC control signal based on the secondary DC output voltage to the secondary auxiliary switching element to execute conduction angle control on the secondary auxiliary switching element to stabilize the secondary DC output voltage, wherein the insulating converter transformer has a core that is not provided with any gap for prohibiting saturation, the primary winding and the secondary winding are wound around the core in the opposite winding directions and the primary winding and the secondary winding are connected to each other so that additive polarity is established.
According to the present invention, there is achieved a so-called composite resonance type switching converter construction in which the primary parallel resonance circuit forming the voltage resonance converter is equipped at the primary side, and the secondary parallel resonance circuit constructed by the secondary winding and the secondary parallel resonance capacitor is equipped at the secondary side. Further, the active clamp circuit is provided at the secondary side, and the voltage stabilizing control is carried out by subjecting the auxiliary switching element of the active clamp circuit to conduction angle control.
On the basis of the above construction, the primary winding and the secondary winding are wound in the insulating converter transformer so that the winding directions thereof are opposite to each other, and the additive polarity connection is carried out on the primary winding and the secondary winding. Accordingly, the magnetic fluxes achieved by the primary winding and the secondary winding act to offset each other, so that the magnetic flux occurring in the core can be reduced and thus the shift to the saturation state can be suppressed. Therefore, the core of the insulating converter transformer in the switching power supply circuit of the present invention is not equipped with any gap which is formed to suppress the saturation.